The present invention relates to a drive circuit for a voltage-controlled switching device, such as insulated-gate bipolar transistor (IGBT), which is used in a power-conversion device, particularly in an inverter for motor speed control, that is, a circuit for accepting external ON/OFF signals as inputs to generate and apply signals for ON/OFF drive directly applied to a control terminal of the voltage-controlled semiconductor switching device. In particular, the invention relates to a drive circuit having a function of minimizing a turn-on time, power dissipation and noise generation in a driven voltage-controlled semiconductor switching element.
In the figures showing below, the same reference signs denote identical or equivalent parts.
FIG. 4 shows an example of a structure of the conventional simplest drive circuit for driving a power switching device, in this case IGBT 1. In this embodiment, means for charging and turning on the IGBT 1 is a P-channel MOSFET 2, and means for turning off the IGBT 1 is an N-channel MOSFET 4.
If an ON signal is input to an ON/OFF-signal input terminal 8, a predriver 7 first provides an output 7b with a low potential (hereinafter also referred to as xe2x80x9cLxe2x80x9d) to turn off the N-channel MOSFET 4, and then provides an output 7a with xe2x80x9cLxe2x80x9d to turn on the P-channel MOSFET 2. When the MOSFET 2 is turned on, it charges the gate capacitance of the IGBT 1 to turn the IGBT 1 ON. This difference in time between the changes in the outputs 7b and 7a is provided in order to prevent a short-circuit current from flowing when both MOSFETs 2 and 4 are simultaneously turned on due to MOSFET gate delay.
On the other hand, if an OFF signal is input to the ON/OFF-signal input terminal 8, the predriver 7 first provides the output 7a with a high potential (hereinafter also referred to as xe2x80x9cHxe2x80x9d) to turn off the P-channel MOSFET 2, and then provides the output 7b with xe2x80x9cHxe2x80x9d to turn on the N-channel MOSFET 4. Thus, the IGBT 1 gate capacitance is discharged and the IGBT 1 is turned OFF.
In driving the power switching devices, such as IGBTs or MOSFETs, it is important to reduce overall losses including steady-state losses caused by an ON voltage, and switching losses, i.e. turn-on and turn-off losses. If, however, the switching speed is increased to reduce the switching losses, rapid dV/dt or di/dt may cause noise. This noise is normally large when the power switching device is turned on.
In recent years, a great effort has been made in developing drive methods that solve this problem in order to reduce both switching losses and noise. Such drive methods are disclosed in Japanese Patent Application Laid Open (KOKAI) Nos. 61-237513 and 7-240676, wherein a capacitor is connected between the gate and drain of a field-effect transistor (FET) to reduce the ON/OFF speed.
FIG. 5 shows an inverter circuit that uses a pulse-width modulation (PWM) for a motor speed control. The circuit has been simplified to facilitate explanation of the operation of a semiconductor switching device. FIG. 6 shows the turn-on waveforms of the IGBT 1 in this circuit. An operation is described with reference to FIGS. 5 and 6.
First, when the IGBT 1 is turned on, a current flows through a load inductance L. The current increases in terms of di/dt (=Ed/L) determined by the inductance L and a power-supply voltage Ed (specifically, Edxe2x88x92(ON voltage of the IGBT 1)=voltage applied to the inductance L).
Next, the IGBT 1 is turned off when a fixed current is reached, and the current flow is commuted through a free-wheeling diode 31 (transition from I0 to I1 in FIG. 5).
Next, when the IGBT 1 is turned on again, all currents flowing through the free-wheeling diode 31 shift to the IGBT 1 at the time that the current through the IGBT 1 has increased to the value of the current through the inductance L (I1 in FIG. 5: in fact, this current is slightly less provided by an ON voltage at the free-wheeling diode 31 or the like, but this attenuation is weak since the inductance L is sufficiently large).
FIG. 6 shows the waveforms of a gate voltage VGE, a collector current Ic, and a collector-emitter voltage VCE observed when the IGBT 1 is turned on under the conditions noted above.
The IGBT 1 is assumed to be driven by the drive circuit in FIG. 4, in which the P-channel MOSFET 2 is turned on at time t1 to start to provide a source current to the gate of the IGBT 1. When the IGBT gate voltage VGE reaches the gate threshold value at time t2, collector current Ic starts flowing and increases with the increase of the gate voltage.
When the collector current Ic comes to equal to the forward current through the free-wheeling diode 31 at time t3, current no longer flows from the inductance L to the IGBT 1, because the inductance L is sufficiently large to thereby suppress rapid increase in current flowing through the inductance L.
However, even when no current is flowing through the free-wheeling diode 31, excess carriers that have been generated by the conductivity modulation while current is flowing remain. If the gate voltage of the IGBT 1 is sufficient to allow a higher collector current Ic to flow, a current (hereinafter referred to as a xe2x80x9creverse recovery currentxe2x80x9d) I2 flows transitionally in a direction opposite to a direction where the current was previously flowing through the free-wheeling diode 31.
This reverse recovery current I2 flows through the IGBT 1 as shown in FIG. 5 such that a current flowing through the IGBT 1 is expressed by I1 (=I0)+I2. The gate voltage VGE of the IGBT 1 attempts to rise continuously due to a charging current from the P-channel MOSFET 2, but falls steadily along with the collector-emitter voltage VCE of the IGBT 1. This is due to the fact that this voltage drop causes current to flow through a collector-gate capacitance of the IGBT 1 (what is known as the xe2x80x9cMiller effectxe2x80x9d).
The reason that the collector-emitter voltage VCE of the IGBT 1 starts to decline (t4) after the reverse recovery current I2 starts to flow through the free-wheeling diode 31 (t3) is now explained. In an area in which the free-wheeling diode 31 can supply a current appropriate for the gate voltage of the IGBT 1 based on a diffusion current associated with a change in an internal carrier distribution, a depletion layer need not extend and the voltage at the free-wheeling diode 31 does not rise, thereby preventing a decrease in the collector-emitter voltage VCE of the IGBT 1.
In an area in which a current can not be supplied unless the reverse recovery current continues to flow through the free-wheeling diode 31 to extend the depletion layer, the reverse recovery current flowing through the free-wheeling diode 31 increases and the collector-emitter voltage VCE of the IGBT 1 starts to decrease. Thus, due to the Miller effect, the gate voltage VGE of the IGBT 1 falls, and the collector current Ic flowing through the IGBT 1 takes a peak value near time t4 and subsequently falls to the value of the current Ii previously flowing through the free-wheeling diode 31. Subsequently, between time t5 and t6, the gate voltage VGE remains at or near the value at which the IGBT 1 can maintain the current previously flowing through the free-wheeling diode 31.
On the other hand, the collector-emitter voltage VCE decreases consistently with dV/dt, as shown in FIG. 6. This is due to the fact that, as the VCE diminishes, the current flowing through the collector-gate capacitance of IGBT 1 is balanced by the charging current from the P-channel MOSFET 2 so that they are equal. This also corresponds to a decrease in the collector-emitter voltage VCE, which narrows the depletion layer in IGBT 1 and so increases the collector-gate capacitance of the IGBT 1.
In FIG. 6, after time t5, the collector current Ic remains constant after the gate voltage VGE becomes constant. In fact, the collector current increases in terms of di/dt determined from the inductance L and a voltage applied thereto, but may be considered to be substantially constant on the time scale shown in this figure, due to the sufficiently large inductance L.
As described above, a current greater than that flowing through the inductance L between time t3 and t5 flows through the IGBT 1. The portion of this current that exceeds the current previously flowing through the inductance L (this portion corresponds to the reverse recovery current flowing through the free-wheeling diode) causes losses in the free-wheeling diode 31 and increases losses in the IGBT 1.
Furthermore, if the reverse recovery current decreases rapidly, a voltage equal to Lsxc2x7di/dt may occur due to the floating inductance Ls shown in FIG. 5, to apply an excessively large voltage to the free-wheeling diode 31 and an IGBT 35 connected in parallel thereto, thereby destroying the device or causing an improper ignition in the IGBT 35. Even in the absence of such a phenomenon, the collector current Ic may vibrate as shown in FIG. 6, thereby increasing a radiated noise.
Next, conventional methods for reducing a turn-on delay and losses in the IGBT 1, and for restraining errors, such as noise, will be described.
FIG. 7 shows an example of a conventional drive circuit that is intended to achieve the above purpose and differs from FIG. 4 in that it has P-channel MOSFETs 36 and 37 connected thereto instead of the P-channel MOSFET 2, and in that the gate of the P-channel MOSFET 36 is not driven directly by the predriver 7, but by a pulse circuit 38.
FIG. 8 is a timing chart useful for explaining the operation of the circuit shown in FIG. 7. When an ON signal is input to the ON/OFF-signal input terminal 8 (port A) of the predriver 7, the predriver 7 first provides the output 7b (that is, the potential at the gate (point D) of the MOSFET 4) with xe2x80x9cLxe2x80x9d to shut down the MOSFET 4. Next, following a delay td1, the predriver 7 provides the output 7a (that is, the potential at a gate of the MOSFET 37 and an input (point B) of the pulse circuit 38) with xe2x80x9cLxe2x80x9d to turn the MOSFET 37 on. The predriver 7 also switches the potential at a gate (point C) of the MOSFET 36 in such a manner that the output 7a is kept xe2x80x9cLxe2x80x9d for the period td2, and then xe2x80x9cHxe2x80x9d for the period td3, and is further switched back to xe2x80x9cLxe2x80x9d via the pulse circuit 38.
The period td2 corresponds to the period between time t1 and t2 shown in FIG. 6, and the period td3 corresponds to the period between time t2 and t4 shown in FIG. 6. That is, the gate of the IGBT 1 is charged only by the MOSFET 37 during the period between time t2 and t4 (period td3), whereas it is charged by the MOSFETs 36 and 37 during the period between time ti and t2 (period td2) and on and after time t4 (following the period td3). This provides the following effects:
First, due to the use of the MOSFETs 36 and 37 for charging between time t1 and t2, the amount of time required from the input of the ON signal until the gate voltage VGE of the IGBT 1 reaches a threshold value (this amount of time corresponds to the turn-on delay time) can be reduced. This does not reduce switching losses in the IGBT 1, but reduces the delay time before the-actual turn-on of the IGBT 1 (start of conduction) to improve control ability.
Next, since the gate of the IGBT 1 is charged only by the MOSFET 37 between time t2 and t4, the di/dt of the collector current flowing through the IGBT 1 decreases to slightly increase the switching losses in the IGBT 1 during the period between time t2 and t3. Since, however, the overshoot of the gate voltage VGE (peak of the gate voltage VGE between time t3 and t1 shown in FIG. 6) declines to reduce the peak value of the collector current IC while restraining rapid increases in Ic. This, in turn, prevents the voltage surge caused by the floating inductance Ls and radiation noise arising from the current vibration.
On and after time t4, the MOSFETs 36 and 37 are used for charging again, so that the Miller effect reduces the period during which the gate voltage of the IGBT 1 is constant (period between time t5 and t6 shown in FIG. 6), thereby facilitating the decrease in the collector-emitter voltage VCE of the IGBT 1 to lessen the turn-on losses during this period.
Thus, in the IGBT 1 of the circuit shown in FIG. 7, the turn-on losses increase during time t2 and t3 shown in FIG. 6, but can be reduced during the period between time t5 and t6, thereby restraining the voltage surge and noise without increasing switching losses (in this case, turn-on losses).
The period td4 shown in FIG. 8 is formed to prevent simultaneous turn-on of the MOSFET 36 or 37 and the MOSFET 4, as in the period td1.
In a practical inverter circuit for a motor control, pulse-width modulation is used to control an output current from the inverter circuit so as to create a sine waveform. As a result, the current flowing through the IGBT 1 is not constant. Accordingly, the duration of the period between time t2 and t3 shown in FIG. 6 is not constant. Thus, the period during which the MOSFET 36 shown in FIG. 7 is turned off while the MOSFET 37 remains on (this corresponds to the period td3 shown in FIG. 8) must be sufficiently long to accommodate the maximum current.
Thus, in fact, the current for charging the gate capacitance of the IGBT 1 remains low just after time t5 shown in FIG. 6, in which the charging current must be high, and as a result, switching losses can not be sufficiently reduced during the period between time t5 and t6. This circuit has another disadvantage in that an appropriate control can not be provided if the time t1 or t2 shown in FIG. 6 varies due to its temperature characteristics or the like.
Next, a situation will be discussed in which the IGBT 1 is turned on if a load on the IGBT 1 is short-circuited. FIG. 9 shows waveforms of the current and voltage observed when the IGBT 1, which was off, is turned on by short-circuiting the load. Charging of the gate capacitance of the IGBT 1 begins at time t11, and when the gate voltage VGE reaches the gate threshold value at time t12, a current starts to flow through the IGBT 1.
During a load short circuit, if the effects of the floating inductance of wiring or the like are neglected, the voltage applied to the IGBT 1 will be constant. Accordingly, the Miller effect associated with the decrease in the collector voltage VCE of the IGBT 1 shown in FIG. 6 prevents the gate voltage VGE of the IGBT 1 from remaining constant. As a result, the gate voltage VGE continues to rise and reaches a voltage value VCC of a power supply 11 for a driving circuit at time t13.
During the short circuit, however, carriers within the IGBT 1 cause a current to flow from the gate of the IGBT 1 toward the drive circuit, and the impedance of the driving circuit and the inductance of the gate wiring cause a voltage drop, thereby increasing the gate voltage VGE to the same level as, or higher than, the supply voltage VCC of the driving circuit. This phenomenon is significant if a gate resistor (connected between the gate of the IGBT 1 and the drain of the MOSFET 2, not shown in FIG. 4) is provided. Then, following a transition period, the gate voltage VGE reaches a maximum value at time t14, and then returns to the supply voltage VCC at time t15.
On the other hand, the collector current Ic increases along with the gate voltage VGE, and then decreases in proportion to the gate voltage VGE after the latter peaks at time t14. It is due to an increase in the temperature of the IGBT 1 that the collector current Ic falls, even after the gate voltage VGE has reached a constant value.
Recent IGBTs have improved in performances due to their reduced ON voltage or the like, but the IGBTs with higher performances have a higher collector current during a load short circuit and are thus more likely to be destroyed in a short period of time. Thus, current-limiting circuits are commonly used to detect a load short circuit to restrain increases in gate voltage.
If, however, the gate voltage and collector current increase at a high rate, destruction may occur before the current-limiting circuit is activated. If, in the circuit shown in FIG. 4, the normal switching speed of the IGBT 1 is not reduced, since the ON resistance of the MOSFET 2 can not be augmented, the collector current of the IGBT 1 increases at a higher speed to hinder a circuit protection.
In the circuit shown in FIG. 7, the increase in speed is controlled in the initial stage of the increase in the collector current, but the gate of the IGBT 1 is rapidly charged after a fixed period of time. The current level at which a short-circuit current is detected is higher than the normal current level, so that in the circuit shown in FIG. 7, the increase in current from the detection of a load short circuit until the protection circuit starts to operate is greater than that in the circuit shown in FIG. 4. Consequently, a short-circuit protection in this circuit is more difficult than in the circuit shown in FIG. 4.
It is an object of the present invention to provide a drive circuit that solves the problems with the above-described conventional drive circuits.
It is another object of the invention to provide a drive circuit that can reduce turn-on delays and losses while restraining possible noise during a turn-on period of a voltage-controlled semiconductor switching device to be driven.
It is a further object of the invention to provide a drive circuit that can be easily protected during the turn-on period in a load short-circuit state.
To attain the above objects, according to the first aspect of the invention, a drive circuit comprises means for supplying a current to a control terminal (gate or the like) of a voltage-controlled semiconductor switching device to be driven (IGBT 1, sensing IGBT 2, or the like) based on the input of an ON signal (to an ON/OFF-signal input terminal 8 of a predriver 9), and means (N-channel MOSFET 4 or the like) for removing a current from the control terminal of the voltage-controlled semiconductor switching device based on the input of an OFF signal (to the ON/OFF-signal input terminal 8). The current-supplying means has a voltage-controlled transistor (P-channel MOSFET 2 or the like) having its drain connected to the control terminal, and a capacitor (10) connected between the gate and drain of the voltage-controlled transistor.
According to a drive circuit in the second aspect, in the drive circuit according to the first aspect, the current-supplying means has a slow-charging means for relatively slowly charging the gate of the voltage-controlled transistor based on the input of the ON signal.
According to a drive circuit in the third aspect, in the drive circuit according to the second aspect, the slow-charging means comprises a semiconductor switching device (N-channel MOSFET 5) and a resistor (6) connected in series therewith.
According to a drive circuit in the fourth aspect, in the drive circuit according to the second aspect, the slow-charging means comprises a semiconductor switching device (N-channel MOSFET 12) having a low current conductance capacity (small channel width/channel length, and a high ON resistance).
According to a drive circuit in the fifth aspect, in the drive circuit according to the second aspect, the slow-charging means is a constant-current source.
According to a drive circuit in the sixth aspect, in the drive circuit according to any one of the first to fifth aspects, the current-supplying means further includes means (gate voltage detecting circuit 13 and P-channel MOSFET 14) for rapidly charging the control terminal of the voltage-controlled semiconductor switching device during the period between the input of the ON signal and the start of the flow of a current through the voltage-controlled semiconductor switching device to be driven (at time t2).
According to a drive circuit in the seventh aspect, in the drive circuit according to any one of the first to fifth aspects, the current-supplying means further includes means for detecting a commutation completion point (t3) when all currents flowing through a free-wheeling diode (31) connected in series with the voltage-controlled semiconductor switching device to be driven has been commuted to the voltage-controlled semiconductor switching device, and means (control circuit 17, P-channel MOSFET 14) for rapidly charging the control terminal of the voltage-controlled semiconductor switching device during the period between the input of the ON signal and the commutation completion point.
According to a drive circuit in the eighth aspect, in the drive circuit according to the seventh aspect, the means for detecting the commutation completion point has means (sample-and-hold circuit 15) for sampling and holding the last current flowing through the voltage-controlled semiconductor switching device, and current-comparison means (comparator 16) for comparing the sampled and held current value with the present current flowing through the voltage-controlled semiconductor switching device.
According to a drive circuit in the ninth aspect, in the drive circuit according to the eighth aspect, the current comparison means adds a predetermined offset (19) to the compared current value to enable earlier detection of the commutation completion point.
The operation of the present invention will be described below. A capacitor is inserted between the gate and drain of the MOSFET used to charge the control terminal of the semiconductor switching device to be driven (which will be referred to as a xe2x80x9cmain power devicexe2x80x9d hereinafter for the sake of convenience) to turn on the device, so that a larger Miller effect can be obtained if the drain voltage of the MOSFET for charging the control terminal, that is, the voltage at the control terminal of the main power device, varies during the turn-on period of the main power device.
Consequently, in an area in which the voltage at the control terminal of the main power device rises (or falls) rapidly, the ON resistance of the control-terminal-charging MOSFET decreases at a reduced (or increased) speed. The speed at which the voltage at the control terminal of the main power device varies, and thus di/dt of a current flowing through the main power supply between time t2 and t5 shown in FIG. 6 and dV/dt measured at the beginning of a decline in the collector-emitter voltage, are maintained at small values to restrain a current peak and thus noise. Furthermore, the speed at which the current increases is reduced during the turn-on period if a load on the main power device is short circuited, thereby allowing the protection circuit to protect the main power device easily.
On the contrary, in an area in which the voltage at the control terminal of the main power device varies slowly, the Miller effect does not operate and the ON resistance of the control-terminal-charging MOSFET falls at an increased rate to shorten the period between time t5 and t6 shown in FIG. 6, in order to lessen the losses during the latter half of the turn-on period (first aspect).
In addition, means for relatively slowly charging the gate of the control-terminal-charging MOSFET is provided to improve the effect of reducing the di/dt and dV/dt of the main power device (second to fifth aspects).
To prevent a delay at the time of turn-on when the above-described method is used, the means is separately provided for rapidly charging the control terminal of the main power device during the period between the input of the ON signal and the start of the flow of current through the main power device (sixth aspect).
Additionally, to enable the above-described method to be used to prevent the former half (period between time t2 and t3 shown in FIG. 6) of the turn-on period from extending to increase losses during this period, the means is separately provided for rapidly charging the control terminal of the main power device during the period between the input of the ON signal and time t3 shown in FIG. 6 (commutation completion time) when all currents flowing through the free-wheeling diode connected in series with the main power device have been commuted to the main power device (seventh aspect).
The means for sampling and holding the last current flowing through the main power device and the current-comparison means for comparing the sampled and held current value with the present current flowing through the voltage-controlled semiconductor switching device are provided as means for detecting the commutation completion time (eighth aspect).
The offset is added to the current-comparison means to prevent a delay in completion of the rapid charging of the control terminal of the main power device, based on the means for detecting the commutation completion time or other means (ninth aspect).